1. Field of the Invention
This invention relates to the conversion of the constituent frequencies of each of a set of baseband signals to one of a set of RF channels such that the baseband signals can be combined and distributed as a single composite broadband signal, and more specifically to an exemplary frequency conversion apparatus used in multichannel systems such as cable TV (CATV) head-ends, necessary for transmission of analog TV signals, digital QAM signals used in digital TV and high speed Internet communications, and more particularly to an improved agile frequency conversion method and apparatus that attenuates broadband system noise using an IF-RF level exchange and attenuates distortion components using tunable notch filters, on a channel-by-channel basis, to a ensure that the aggregate carrier-to-noise (C/N) and carrier-to-distortion (C/D) performance levels specified for the multichannel system are met.
2. Background of the Related Art
In typical broadband multichannel systems such as those used for cable television transmission, a set of baseband television signals are combined for simultaneous transmission and distribution in the form of a single composite broadband signal. For a cable television system, a subscriber receives the entire broadband signal simultaneously and discriminates between (or tunes receiving equipment to) one of the unique channels assigned to each of the base-band signals to view the selected base-band signal. To generate the broadband signal, each of the base-band signals must modulate one of a unique set of RF carrier frequencies, each set assigned to one of the channels so that a subscriber may tune receiving equipment to that RF channel frequency carrying the desired baseband signal. The receiving equipment then demodulates the carrier(s) to recover the baseband signal for viewing and listening.
Typical services offered to subscribers in a modern CATV plant include analog TV, digital services via high-speed data modems for digital TV and high speed Internet communications. The analog TV utilizes analog amplitude modulation, while digital services use quadrature amplitude modulation with 64 or 256 levels (i.e. 64 or 256 QAM). The composite broadband signal is generated at the head-end of the cable TV system for distribution to subscribers. While this composite signal typically includes signals carrying analog modulation combined with signals carrying digital modulation, the manner in which it is generated can be illustrated by the following steps, using analog modulation as an example.
First, each of a set of baseband signals (5, FIG. 3) is modulated on the same standard IF carrier frequency. In the case of video signals, the baseband signal (occupying a frequency range of DC to 4.2 MHz) modulates an IF sub-carrier IFC(vid) 9 of 45.75 MHz using amplitude modulation (AM). Baseband audio signals (occupying a frequency range of 20 Hz to 20 KHz) modulate an IF sub-carrier IFC(aud) 8 of 41.25 MHz using frequency modulation (FM). For each of the baseband signals, these two modulated components are combined to create a composite modulated IF signal (10, FIG. 1), which includes the modulated sub-carrier components 8 and 9 and all of the associated side-band components generated by the modulation process.
To assign each of the baseband signals to a unique RF channel, each of the composite IF signals and their associated modulation side-band components are up-converted to occupy a unique one of a set of contiguous RF channel frequency bands. Although each up-converted signal occupies a range of frequencies within its assigned RF channel band, those of skill in the art will recognize that it is often easier to refer to or illustrate an up-converted signal by a primary one of its up-converted sub-carrier components for simplicity. In this description using analog TV as an example, the video sub-carrier of the various composite signals will often be used as a reference to the entire signal.
A conceptual representation of this up-conversion process is illustrated in FIG. 1. The RF channel frequency bands 12 are typically contiguous, 6 MHz wide and occupy an RF channel frequency range 14 of about 50 MHz to about 870 MHz (i.e. about 137 channels at 6 MHz per channel). The up-converted signals are normalized to a certain amplitude, typically amplified to a desired output level, and then summed together to form the composite broadband signal to be distributed. In this way, several baseband modulated IF signals that would otherwise occupy the same frequency range can be transmitted together without significant interference with one another.
Because the circuitry used to implement the up-conversion process is not perfect, however, interference between channels in the form of noise and distortion can still cause unacceptable performance in such systems. The composite broadband signal contains not only the up-converted baseband modulated IF signals occupying their assigned RF channel frequency ranges, but also distortion components (including harmonics and inter-modulation products) and cumulative broadband noise associated with each of the up-converted IF composite signals (artifacts of the imperfect up-conversion process and apparatus). Attenuating the power levels of these noise and distortion components to reach acceptable levels of aggregate system performance poses one of the more significant and challenging problems faced by designers of such broadband multichannel systems.
FIG. 2 illustrates one of the n RF channel bands 12 of RF channel frequency range 14, along with the up-converted composite IF signal of interest 11 having an RF carrier frequency (RFC) 13, various distortion components 20 and system noise 22 generated during the up-conversion and modulation processes. For a single channel in isolation, these noise and distortion components do not pose a problem because they typically fall outside of the channel band 12 for the signal of interest 11. When 60 to over 130 channels are summed together to create the composite broadband output signal for systems such as CATV, however, the distortion components can be superimposed over, and can therefore interfere with, other composite RF signals occupying other RF channel bands 12 in the RF frequency range 14. Moreover, the system noise 22 generated during the up-conversion of each of the channels is summed together such that the noise floor 24 in the aggregate can rise to unacceptable levels with respect to the levels of the composite RF signals.
FIG. 3 is a conceptual representation of the summing process that occurs to create a composite broadband signal at the head end of a multichannel system such as CATV. Each channel processor 30 receives source signals Sn 5, and generates a frequency up-converted composite signal RFOUT(n) 33 that carries the base-band source signal information and falls within one of the RF channel frequency bands. Each of the RFOUT signals 33 is then summed together using conceptual summing element 32 prior to distribution of the composite broadband signal RFBB 34.
The channel processors 30 are typically designed to normalize the output power levels of the RF signals and amplify them to the desired output level so that they are equal before summing them together. Thus, the distortion components produced by the up-conversion process for any of the channels will all nominally have the same power relative to the signal RFOUT 33, including its RF channel carrier component (i.e. RFC 13, FIG. 2) as well as all of the carrier components of any of the other n channels. It follows then that the aggregate carrier-to-distortion ratio (C/D) permitted for the system can be met by optimizing distortion levels generated by the up-conversion process for channels on an individual basis. With respect to system noise, however, the noise generated by each channel processor is cumulative and thus noise reduction must be optimized for all of the channels in the aggregate.
In today's CATV systems, it is expected that each channel should have no less than a 65 dB attenuation of both distortion components and cumulative broadband noise relative to the signal power of the RF carrier signal components of each of the channels. These two specifications are commonly referred to as aggregate carrier to distortion (C/D) and aggregate carrier to noise (C/N) ratios respectively.
One seemingly obvious approach to solving the problem of sufficiently attenuating distortion and noise signals in the broadband signal is to filter each RF channel output signal RFOUT (33, FIG. 3) through a band-pass filter substantially tuned to the RFC (channel carrier frequency) of the assigned channel to attenuate the out-of-band distortion and noise signals 20 and 22 of FIG. 2. The problem with this approach is that each channel processor 30 would have to be manufactured with a different band-pass filter having a center frequency dictated by the frequency band 12 of the channel to which it is assigned. Thus, each channel processor would be dedicated to a specific channel (or range of channels), rather than having the agility necessary to operate for any one of the RF channels in the system. This type of static design is contrary to today's multichannel systems applications that demand agility in design such that each channel processor 30 is capable of assignment to any channel in the RF channel frequency range (14, FIG. 1).
Moreover, this static solution would require equipment manufacturers to manufacture, test and stock different channel processors 30 for each channel (or range of channels), increasing manufacturing costs and requiring purchasers to maintain an inventory of replacement parts for each of the channel processors 30. Because a cable television system can provide between 60 and over 130 channels, this solution becomes impractical and cost prohibitive. Nor can tunable band-pass filters be used to render this solution agile, because tunable bandpass filters include non-linear elements that would themselves introduce distortion components into the broadband signal, making the 65 dB C/N and C/D specifications virtually impossible to meet or exceed.
FIG. 4 is a conceptual illustration of a known channel processor 30. Those of skill in the art will recognize that channel processors 30 may contain additional elements, but the elements pertinent to this discussion include up-converter 38 and modulator 36. Modulator 36 performs the modulation of a baseband source or information signal S 5 (e.g. video 35 and audio 37 signals for cable television) on one or more IF carrier signals to create the composite signal IF 31. As previously discussed, in a CATV system the IF signal 31 is a composite signal incorporating base-band signals for video and audio each modulating two separate sub-carriers. Composite signal IF 31 is then input to up-converter 38, which up-converts the composite signal IF 31 (including all of its modulation side-band components) to a composite RF output signal RFOUT 33 occupying the RF channel frequency band 12 corresponding to the assigned RF channel.
FIG. 5a provides a simple conceptual representation of a single-stage up-converter 38s. Composite signal IF 31, having at least one sub-carrier frequency component IFC and its associated modulation side-band components, is input to a mixer 45 and combined with a local oscillator signal LO 56 to produce composite signal RF′ 39. The constituents of composite signal RF′ 39 are conceptually illustrated in FIG. 5b, which include upper 51 and lower 42 side-band RF signals, system noise and harmonic distortion components (not shown), and local oscillator and IF leakage components LO(L) 44 and IF(L) 58 respectively. Thus, the result of the mixing process is that the composite signal IF 31, including all of its associated modulation side-band components, is up-converted to produce upper and lower side-band signals 51 and 42 respectively, and wherein each now comprises an up-converted RF channel carrier component. Channel sub-carrier frequencies RFC(upp) 53 and RFC(low) 41 are related to IFC 9 (FIG. 1) by the following equations: RFC(upp)=LO+IFC and RFC(low)=LO−IFC. Those of skill in the art will recognize that the other frequency components of composite signal IF 31 have also been up-converted to a frequency equal to their original IF frequency plus and minus LO. Hereafter the composite signals typically will be referred to by one of their primary sub-carrier frequency components for simplicity.
As illustrated in FIG. 5b, because of leakage through the imperfect mixer 45, an oscillator leakage component LO(L) 44 is produced at the frequency of LO as well as a leakage component IF(L) 58 at the primary carrier frequency IFC 9 of the composite signal IF 31. Composite signal RF′ 39 is passed through bandpass filter 54 having transfer function 48 to produce a single side-band composite RF output signal 42, while eliminating the upper side-band and leakage components. Up-converted RF output signal 42 is then conditioned by circuit block 59 to normalize the signal output levels and typically to amplify the signal levels of RFout 33, before combining it with the other up-converted channel outputs to produce the broadband signal RFBB 34 for transmission/distribution.
To isolate the desired side-band carrier component 42 of RF′ 39 and to meet the aggregate performance specifications required of a multichannel system such as cable television, the levels of signal components 58, 44 and 51 of RF′ 39 must be generated at or attenuated to a level that is at least 65 dB below the signal level of carrier component RFC(low) 41. The signal level of carrier component 41 starts out already 7 dB below the signal level of IFC 9 of IF 31 because of conversion loss of about 7 dB associated with the mixer 45. Moreover, mixer 45 can require a range of power levels for the oscillator signal LO 56 on the order of 7 dBm to 21 dBm. Although it might be desirable to overcome the conversion loss of the mixer by increasing the power of IF 31, this will cause the levels of distortion components to increase on RF′ 39. Thus the upper limit to the input level of IF 31 is approximately −10 dBm and the output power of RF′ 39 will be −17 dBm. If the mixer 52 requires the signal LO 56 to be 20 dBm and the LO rejection is approximately 25 dB, then the oscillator leakage component LO(L) 44 will be at about −5 dBm, and therefore approximately 12 dB hotter than the side-band carrier signal component RFC(low) 41. Therefore, to reach the −65 dB specification, a filter must actually attenuate the oscillator leakage component LO(L) 44 by at least 77 dB. Such a response is difficult to achieve even with a fixed band-pass filter let alone one that is tunable.
In an attempt to meet this difficult performance specification while maintaining an agile system, a dual or two-stage frequency conversion has been employed. FIG. 6a illustrates the concept of a prior art dual or two-stage frequency up-converter 38d. For the first conversion stage, modulated composite signal IF 31 is input to mixer 52 along with a local oscillator signal LO1 66 and a resulting composite signal IF′ 60 is produced and input to a fixed bandpass filter 64. Band-pass filter 64 then produces composite signal IF1 68. This first conversion stage operates in the same manner as the single conversion circuit 38s of FIG. 5a, producing an up-converted IF output signal IF′ 60 analogous to RF′ 39 illustrated in FIG. 5b. The primary difference is that the first stage up-converts the modulated composite signal IF 31 to intermediate composite signal IF1 68 having a carrier frequency much greater than the RF channel frequency range 14 of the system. For the second conversion stage, IF1 68 is input to a second mixer 61 and mixed with a second local oscillator signal LO2 63 to produce composite RF signal RF′ 65. The frequency of oscillator signal LO2 63 is chosen such that the lower side-band component of RF2 65 falls within the appropriate channel band 12 of the RF channel frequency range 14, corresponding to the channel to which the channel processor 30 is currently assigned. RF′ 65 is then input to RF attenuator 67, which produces an output RFN 57 that is normalized to a constant power level relative to each of the RF outputs of the other channel processors 30 of the system.
FIG. 6b provides a conceptual illustration of the constituent signals produced by this dual conversion process. As previously discussed, the first conversion stage up-converts its composite modulated signal IF 31 to produce output IF′ 60, which includes upper and lower side-band components 71 and 74 respectively. For a CATV system, the frequency of oscillator signal LO1 66 is chosen such that the frequency of the primary carrier IF′C(Upp) 79 of upper side-band component 71 is equal to approximately 1 GHz; the frequency of IF′C(Upp) 79 is equal to LO1+IF. It should be noted that the oscillator leakage component LO1(L) 73 corresponding to local oscillator signal LO1 66, as well as lower side-band component 74, fall outside of the RF channel frequency range 14. Lower side-band component 74, as well as local oscillator leakage component 73 are then attenuated by fixed bandpass filter 64 having transfer function 72 such that composite signal IF1 68 contains only upper side-band component 71.
The second conversion stage then down-converts upper side-band component 71 of IF1 68 to a one of a range of frequencies that falls within the RF channel band 12 of RF channel range 14 that corresponds to the channel to which channel processor 30 is currently assigned. The down-conversion of composite signal IF1 68 is effected by mixing IF1 68 with local oscillator signal LO2 63 using mixer 61 to produce a second converted composite signal RF′ 65, which includes lower side-band component 76, an upper side-band component (not shown), and oscillator leakage component LO2(L) 75(L–U). The frequency of local oscillator signal LO2 63 is chosen such that the primary RF sub-carrier frequency component RF′C 62 of the lower side-band component 76 of composite signal RF′ 65 falls within the assigned RF channel frequency band 12. The frequency of lower sideband carrier component RF′C 62 is equal to LO2−IF1C, where IF1C is the primary sub-carrier component of IF1 68.
FIG. 6b also illustrates the range of operation of the dual conversion for an exemplary cable TV system. For each channel processor 30, LO2 63 has one of a set of frequency values falling within the range of about 1070 MHz to about 1880 MHz (corresponding to a range of oscillator leakage components 75L–U. Each value of LO2 63 produces a lower side-band component 76L–U of RF′ 65 having a frequency range falling within one of the set of RF channel bands 12 ranging from RF channels 1 (12L) through n (12U). Note that the upper sideband component of RF′ 65 (not shown) having a primary sub-carrier frequency equal to LO2+IF1C, and the range of oscillator leakage components LO2(L) 75L–U associated with local oscillator signal LO2 63, fall outside of the RF channel frequency range 14 for the entire range of the system's operation.
Those with skill in the art will recognize that this prior art two-stage conversion technique provides system agility by eliminating the need to attenuate leakage and unwanted side-band components from the composite signal RF′ 65 using filtering techniques. No filtering of these components is required because they all fall outside of the RF channel frequency range 14. Thus, each channel processor 30 can be assigned to up-convert composite IF signals to any one of the RF channels contained in channel range 14 simply by adjusting the value of local oscillator frequency LO2 63 between the frequency values of about 1070 MHz and approximately 1880 MHz.
Eliminating the need to filter these components does not, however, solve all of the problems associated with higher-order distortion components and broadband noise. With respect to distortion, two additional second-order distortion components generated by the prior art two-stage conversion above still find their way into the RF channel range and become the dominant limiting factor on distortion performance in the up-converters of FIGS. 5a and 6a. One such component is the second harmonic 2RF′ 85 of the composite signal RF′ 65, which has a sub-carrier frequency component 81 having a frequency equal to 2RF′ as illustrated in FIGS. 7a and 7b. A second distortion component 86 of RF′ 65 has a primary carrier component 82 the frequency of which is equal to IFC−RF′C. As can be seen from FIGS. 7a and 7b, as the frequency of RF′C 62 increases, the second harmonic component 81 increases in frequency at the rate of 2RF′C and distortion component 82 decreases in frequency and moves right to left on the frequency graph of FIGS. 7a and 7b. 
Second harmonic distortion sub-carrier component 81 presents a problem for lower channels in a CATV system because for the lower half of the channel frequency range 14, the second harmonic component given by 2RF′ still falls inside the channel frequency range 14, and thus interferes with other channels. On the positive side, component 81 is always filterable because it is always at a frequency of twice the frequency of the sub-carrier of the signal of interest, RF′C 62. As can be seen from FIG. 7b, however, distortion sub-carrier component 82 (it moves from right to left with increasing frequency of RF′C 62), can become too proximate in frequency to RF′C 62 such that it cannot be filtered without attenuating the up-converted signal of interest RF′ 65 as well. Component 82 can for some frequencies even fall directly within the very channel frequency band 12 to which the up-converted composite signal RF2 65 is assigned.
The typical prior art approach to addressing the attenuation of second-order distortion components 81, 82 is to simply design all of the channel processors 30 such that the power output levels of the distortion components 82, 81 will always be at least 65 dB down from the power level of RF′C 62. If so designed, no attenuation of distortion components 81, 82 by filtering is required. This is accomplished through the design of mixer 61 and the choice of input levels for signals IF1 68 and LO2 63. Put another way, mixer 61 (FIG. 6a) must be constrained by design and operation such that the distortion components 81, 82 (FIGS. 7a and 7b) are guaranteed to always be a minimum of 65 dB down from the power level of the carrier signal component RF′C 62 of RF′ 65. Regrettably, this technique for meeting the C/D specification comes at the expense of the C/N ratio as explained below.
To meet the C/D specification, the primary constraint on the operation of mixer 61 will be the maximum input level of IF1 68 into mixer 61. For every 1 dB increase in signal power of IF1 68, the distortion components 81, 82 (the frequencies of which are equal to 2RF′C and IF1C−RF′C respectively) will increase by 2 dB. Thus, for every 1 dB of increase in IF1 68 there will be a loss of 1 dBc/dB in the C/D ratio. As a result, an upper limit must be imposed on the input level that is permissible for IF1 68 to ensure compliance with the C/D specification. The problem with constraining the input level of IF1 68, however, is that the output level of RF′ 65 will also be lower with respect to the broadband system noise floor, thereby lowering the C/N performance. This problem is further exacerbated by the fact that the lower output level of RF′ 65 typically must be compensated through post-mixing amplification, using amplifier 55, which not only amplifies the output level of RF′ 65, but also the system noise level at the RFoutput 33.
Thus, the prior art two-stage frequency converter of FIG. 6a must still somehow meet the cumulative broadband or system noise performance requirement of −65 dB, despite the constraints being placed on the pre-mixer input level to mixer 61. The power level or noise floor 24 of the system noise for signal RF′ 65 is independent of the input levels of mixer 61. As previously discussed, however, the C/N ratio decreases as the input level to the mixer is decreased. Moreover, the absolute noise floor increases with the post-mixer gain (achieved using amplifier 55) required by the lower output level. Because the broadband noise of a broadband composite signal is cumulative as the signals are summed, the noise level for a single channel is multiplied by the number of channels n, making it extremely difficult to meet the aggregate C/N ratio for RFBB 34 as specified for the system.
To meet the aggregate C/N specification, prior art systems often employ a switched bank 69 of m switched band-pass filters 53 that essentially divides the RF channel frequency range 14 into m frequency ranges, each encompassing multiple contiguous channels within their respective pass-bands. The typical number of filters used is between six and eight to divide the RF channel frequency range 14 into sub-octave groups of channels. The channel processor 30 selects the appropriate one of the six to eight filters 53 designed to pass a range of frequencies that encompasses the frequency band 12 to which the channel processor is assigned. Thus, each of the channel processors 30 must switchably select, in accordance with its assigned frequency channel, a final band-pass filter 53 through which the normalized RFN signal 57 is filtered to attenuate broadband noise falling outside of the pass-band of the filter. In this way, when the signals are summed to form RFBB 34 of FIG. 3, the channels falling inside of the pass-band of the filter will receive significantly attenuated noise contributions from the channels falling outside of the pass-band. Thus, when all of the channels of the system are combined into composite signal RFBB 34, the cumulative broadband noise will fall below the aggregate C/N performance specification of −65 dB for the system.
This solution is expensive because to maintain agility, each of the channel processors 30 must have all of the switched band-pass filters and must also have the mechanism by which to select and switch in the appropriate filter, in accordance with the processor's channel assignment. Moreover, as the performance level demanded by multichannel systems continues to increase, it will be more difficult for these switched band-pass filters to provide sufficient attenuation by which to meet the composite noise performance specifications. Additionally, there must be sufficient amplification of the RFN output signal 57 not only to overcome the lower input levels as previously discussed, but also to overcome the insertion losses associated with the bank of band-pass filters 69. Finally, such an amplifier must be able to accomplish this amplification with low distortion, which is expensive and consumes a considerable amount of power.
A prior art implementation of an amplifier (55, FIG. 6a) typically employed in this context is illustrated in FIG. 14. The amplifier employs a well-known push-pull topology that cancels out second-order distortion terms generated by amplifiers 400 and 402.
Thus, there is room in the art for an agile method and apparatus for up-converting modulated base-band signals to RF frequencies to produce a broadband composite signal that sufficiently attenuates undesirable distortion components in the composite signal to meet and preferably exceed the aggregate C/D specification for the system, while reducing the broadband noise of the composite signal to meet and preferably to exceed the aggregate system C/N specification without the need for large numbers of costly switched filters and power-hungry amplifiers.